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Senin, 04 Januari 2010

Guitar Control

Stand-alone, 9V battery powered unit

Three-level input selector, three-band tone control


P1,P2_________100K Linear Potentiometers
P3____________470K Linear Potentiometer
P4_____________10K Log. Potentiometer

R1____________150K 1/4W Resistor
R2____________220K 1/4W Resistor
R3_____________56K 1/4W Resistor
R4____________470K 1/4W Resistor
R5,R6,R7_______12K 1/4W Resistors
R8,R9___________3K9 1/4W Resistors
R10,R11_________1K8 1/4W Resistors
R12,R13________22K 1/4W Resistors

C1____________220nF 63V Polyester Capacitor
C2,C8___________4µ7 63V Electrolytic Capacitors
C3_____________47nF 63V Polyester Capacitor
C4,C6___________4n7 63V Polyester Capacitors
C5_____________22nF 63V Polyester Capacitor
C7,C9_________100µF 25V Electrolytic Capacitors

IC1___________TL062 Low current BIFET Dual Op-Amp

J1,J2__________6.3mm. Mono Jack sockets

SW1______________1 pole 3 ways rotary or slider switch
SW2______________SPST Switch

B1_______________9V PP3 Battery

Clip for PP3 Battery

Device purpose:

This preamplifier was designed as a stand-alone portable unit, useful to control the signals generated by guitar pick-ups, particularly the contact "bug" types applied to acoustic instruments. Obviously it can be used with any type of instrument and pick-up.
It features a -10dB, 0dB and +10dB pre-set input selector to adjust input sensitivity, in order to cope with almost any pick-up type and model.
A very long battery life is ensured by the incredibly low current consumption of this circuit, i.e. less than 800µA.
Circuit operation:

IC1A op-amp is wired as an inverting amplifier, having its gain set by a three ways switch inserting different value resistors in parallel to R4. This input stage is followed by an active three-band tone control stage having unity gain when controls are set in their center position and built around IC1B.
Technical data:

Frequency response:20Hz to 20KHz -0.5dB, controls flat.

Tone control frequency range: ±15dB @ 30Hz; ±19dB @ 1KHz; ±16dB @ 10KHz.

Maximum input voltage (controls flat): 900mV RMS @ +10dB input gain; 7.5V RMS @ -10dB input gain.

Maximum undistorted output voltage: 2.5V RMS.

Total Harmonic Distortion measured @ 2V RMS output: <0.012% @ 1KHz; <0.03% @ 10KHz.

THD @ 1V RMS output: <0.01%

Total current drawing: <800µa.

Sabtu, 02 Januari 2010

4 Digit Keypad Switch

This is a universal version of the Four-Digit Alarm Keypad. I've modified the design of the output section - to free up the relay contacts. This allows the circuit to operate as a general-purpose switch. I used a SPCO/SPDT relay - but you can use a multi-pole relay if it suits your application.

Do not use the "on-board" relay to switch mains voltage. The board's layout does not offer sufficient isolation between the relay contacts and the low-voltage components. If you want to switch mains voltage - mount a suitably rated relay somewhere safe - Away From The Board.

Schematic Diagram:


The relay is energized by pressing a single key. Choose the key you want to use - and connect it to terminal "E". Choose the four keys you want to use to de-energize the relay - and connect them to "A B C & D". Wire the common to R1 and all the remaining keys to "F".

The Circuit is easy to use. When you press "E" - current through D2 & R9 turns Q6 on - and energizes the relay. The two transistors - Q5 & Q6 - form a "Complementary Latch". So - when you release the key - the relay will remain energized.

To de-energize the relay - you need to press keys "A B C & D" in the right order. When you do so - pin 10 of the IC goes high - and it turns Q4 on through R8. Q4 connects the base of Q6 to ground. This unlatches the complementary pair - and the relay drops out.

Any keys not wired to "A B C D & E" are connected to the base of Q3 by R7. Whenever one of these "Wrong" keys is pressed - Q3 takes pin 1 low and the code entry sequence fails. If "C" or "D" is pressed out of sequence - Q1 or Q2 will also take pin 1 low - with the same result. If you make a mistake while entering the code - simply start again.

The Keypad must be the kind with a common terminal and a separate connection for each key. On a 12-key pad - look for 13 terminals. The MATRIX TYPE with 7 or 8 terminals WILL NOT WORK. With a 12-key pad - over 10 000 different codes are available. If you need a more secure code - use a bigger keypad with more "Wrong" keys wired to "F". A 16-key pad gives over 40 000 different codes.

The Support Material for this circuit includes a step-by-step guide to the construction of the circuit board, a parts list, a detailed circuit description and more.

Expandable Graphic Equaliser


The project described in this article is a constant Q, fully expandable graphic equaliser. Where most "conventional" graphic EQ circuits have a Q that is dependent on the setting of the pot, this one maintains the same Q at all settings. This is achieved by using MFB (Multiple Feedback Bandpass) filters, instead of the more common "gyrator" tuned circuit.

As always, there are pros and cons for the approach described here. Phase shifts tend to be a little more radical, and the passband has more ripple than a conventional circuit, but only where a number of sliders are set to boost or cut. On the positive side, specific frequencies are dealt with specifically regardless of the level, and not with a variable Q. The constant Q circuit makes room equalisation and feedback reduction far better behaved.

So much better in fact, that a boost or cut of 3dB or less may provide the required effect, where a variable Q equaliser may need considerably more, and will affect the adjacent frequencies to a far greater degree.


The filters used are the same as in the Instrument Graphic Equaliser and subwoofer equaliser (see Project 64 and Project 84), and are multiple feedback bandpass types. An example of this filter is shown in Figure 1, and more details are available from the project page for the MFB filter (Project 63). Depending on the configuration you ultimately decide upon, you will need between 10 and 30 of these filters - per channel for stereo!

Figure 1 - Basic Multiple Feedback Bandpass Filter

This circuit is reproduced from the original article for convenience - the actual filter circuits used are slightly different, and are shown in Figure 3. Building 60 of these may sound like an awful chore, which is perfectly reasonable, since it will be just that. With this knowledge at hand, this may go some way to help you make some ...

Now you have to decide on the frequency resolution. 1/3 octave would be really nice, but the number of sliders can be a nightmare. At the very least, you will need octave band, and the suggested (and industry standard) frequencies are ...

3163125250500 1k02k04k08k016k
Octave Band Frequencies

Should you decide on 1/2 octave band frequencies, 20 sliders will cover the range suggested (plus a bit) - these might be ...

31446387125 175250350500700 1k01k42k02k84k0 5k68k011k16k20k
1/2 Octave Band Frequencies

Lastly, 1/3 octave band needs 30 sliders to cover the full frequency range, but the 25Hz and 20kHz bands may not be needed. This still requires 28 slide pots, but the flexibility is greater than you will ever get with conventional tone controls ...

3140506380 100125160200250 315400500630800 1k01k21k62k02k5 3k24k05k06k38k0 10k12k16k
Octave Band Frequencies

There is no reason at all that the unit has to be 1/2 octave or 1/3 octave all the way. The midrange can be 1/3 octave for finest control, but go to 1/2 octave at the extremes. Especially for guitar and bass, I would prefer 1/3 octave up to 1kHz, then 1/2 octave from 1kHz to 8kHz. The final slider would be a 1 octave band filter at 16kHz. The sequence now looks like this ...

3140506380 100125160200250 315400500630800 1k01k42k02k84k0 5k68k016k
Variable Octave Band Frequencies

This gives 23 filters and slide pots, a reasonable compromise that should give excellent results. To ensure reasonable continuity, the filters at 1kHz and 8kHz will need to be a compromise. 1/3 octave filters need a Q of 4, and 1/2 octave filters use a Q of 3, so the 1kHz filter will actually have a Q of 3, and the 8kHz filter will be best with a Q of 2. This might look daunting, but the MFB Filter design program will make short work of determining the component values. Unfortunately, this is only available for users of Microsoft Windows. If you want to use the frequencies shown above, the following table shows the values for each filter.

FreqR1R2R3C1, C2
FreqR1R2R3C1, C2

I have tried to keep the values reasonably sensible. This is not easy with 1/3 octave band equalisers, but all in all the results are quite acceptable (not too many different values). Note that the Q of the filters is changed as the frequency increases - feel free to use the calculator to reverse calculate the values to see the actual gain, Q and frequency error. None of these will be significant in use.

Input / Output Stage

The heart of the circuit is shown in Figure 2. It is not complex, but care is needed to make sure that the opamps do not oscillate. Supply bypassing is critical, and 100nF ceramic caps must be used between supply pins at each opamp package.

Figure 2 - Input / Output Stage

There is one thing of special note in this circuit. R6 (39k as shown) determines the maximum amount of boost and cut, and if you wanted to, you can make it variable. With the filter circuits shown below, 39k allows a boost and cut of 12dB - which is about right in most installations. A value of 10k will allow a maximum of a little over 5dB. Any value between these limits will provide the optimum for a given environment, and this can be preset. This is a very useful feature, and one that I believe is unique to this circuit.


Determining the required Q is the first step in the design process. The requirements are shown in the following table. The gain in each case is unity (actually -1, meaning a gain of unity, but the signal is inverted, or 180 degrees out of phase).

BandwidthRequired Q
1/3 octave4
1/2 octave3
1 octave2

The filters are all connected in the same way, and I do not intend to draw all 30 of them! Instead, I shall show two complete and two partial filters - you will be able to take it from there. The tables above, and/ or the MFB calculator program can be used to determine the values for each individual filter.

Figure 3 - The Filter Bank (Partial Only)

The slide pots are wired with all the end connections in parallel, and the "Sig" output above must drive all the filter inputs, which are also in parallel. For a 1/3 octave equaliser, this represents a load of around 800 ohms on U2B. The NE5532 was chosen as it is one of the few opamps that will drive such a low impedance. Don't be tempted to use anything that is not rated to drive such a low impedance, or it will distort because of output current limiting. Another suitable opamp is the OPA2134 (dual), which also has a very high drive capability - there are no doubt others, but these are the ones I know about.

The maximum rated input voltage is 1V (0dBu), and if you anticipate that the input will be higher than that, I suggest an attenuator at the input. The gain can be restored by increasing the value of R8, so if a 3:1 attenuator were used at the input (10dB), then a 30k (33k would be OK) resistor in place of the 10k will bring the overall gain back to unity.

Remember that U2B operates with gain (about 12dB), so the internal overload limit is lower than you might expect. Because of the narrow bandwidth of each filter, these too can be driven into clipping if the input level is too high, and this is unlikely to improve the sound.

Overall, this is a very versatile unit, and once the initial shock of construction has passed, can be used for the most demanding of equalisation tasks. It can also be used in an automotive installation, but an artifical earth must be created, and the signal voltage limits will be reduced considerably. I suggest that the maximum input voltage be kept below 0.5V RMS - lower than this will provide a better safety margin, and will ensure that clipping does not occur regardless of slider settings.

Figure 4 - Frequency Response of a Single Filter

Figure 4 shows the boost and cut of a single filter, centred on 100Hz. This clearly shows that the Q remains constant - a conventional graphic EQ would have a very broad peak at lower settings, so broad in fact that it would show some noticable effect even at the frequency extremes. Assuming that the 50% pot setting is flat, these graphs were taken at 35/65%, and 0/100% of the pot travel (cut/boost respectively).

This was generated using a 39k resistor for R6 in the input circuit - lower values reduce the maximum boost and cut, but leave the Q unchanged.

Better Volume (and Balance) Controls

Better Volume Control (Pt 1)

The volume control in a hi-fi amp or preamp (or any other audio device, for that matter), is a truly simple concept, right? Wrong. In order to get a smooth increase in level, the potentiometer (pot) must be logarithmic to match the non-linear characteristics of our hearing. A linear pot used for volume is quite unsatisfactory.

Unless you pay serious money, the standard "log" pot you buy from electronics shops is not log at all, but is comprised of two linear sections, each with a different resistance gradient. The theory is that between the two they will make a curve which is "close enough" to log (or audio) taper. As many will have found out, this is rarely the case, and a pronounced 'discontinuity' is often apparent as the control is rotated.

Figure 1 - Circuit of the Log Pot Approximation Figure 2 - The Transfer Curve in dB

Take a 100k linear pot (VOL), and connect a 15k resistor (R) as shown above, to achieve the curve shown. It should be a straight line, but is actually still far more logarithmic than a standard log pot. For stereo, use a dual-gang pot and treat both sections the same way. Use of a 1% resistor for R is recommended. Different values can be used for the pot, but keep the ratio between 6:1 to 10:1 between the value of VOL and R respectively. While 6.7:1 is close to a real log curve, it may still allow excessive sensitivity at low levels. Higher ratios than 10:1 can be used, but will cause excessive loading of the driving stage, or necessitate the use of a pot whose resistance is too high.

As can be seen, provided the gain structure of the preamp is set up properly, a good approximation to true log pot operation is obtained over at least a 25dB range, which is sufficient for the normal variations one requires.

The gain structure of the preamp is correct when the pot spends the vast majority of its time between the 10 and 2 o'clock positions. If the volume is often below or above this range, consider changing the preamp gain.

The other advantage of the 'fake' log pot is that linear pots usually have better tracking (and power handling) than commercial log pots, so there will be less variation in the signal between left and right channels. This is improved even further by the added resistor, which will allow a cheap carbon pot to equal a good quality conductive plastic component (at least for accuracy - I shall not enter the sound quality debate here).

Make sure that the source impedance is low (from a buffer stage) and that it can drive the final impedance when the control is fully advanced (13k Ohms). Use of a high impedance drive will ruin the law of the pot, which will no longer resemble anything useful.

Better Volume Control (Pt. 2 - Further Ideas)

Originally designed by Peter Baxandall (of feedback tone control fame, amongst many other designs), there is also an active version of the 'Better Volume Control', which uses an opamp and a pot in the feedback loop. The log law is almost identical to that for the passive design above, but it can provide gain as well as attenuation. An example of this design may be found in Project 24, and the circuit for the basic idea is shown in Figure 3.

Figure 3 - Active Logarithmic Volume Control

The input buffer enables the inverting stage (needed so the circuit can work) to have a very high input impedance. This would otherwise not be possible without the use of extremely high value resistors, which would increase the noise level considerably. The maximum gain as shown is 10 (20dB) and minimum gain is 0 (maximum attenuation). The input impedance is variable, and is dependent on the pot setting. At minimum gain, input impedance is the full 50k of the pot, falling to about 27k at 50% travel, and around 4k at maximum gain.

These impedance figures are very similar to the simple passive version (if a 100k pot is used), and again, a low impedance drive is required or the logarithmic law will not apply properly. The actual value for VR1 does not matter, and anything from 10k to 100k will work just as well, although it will influence the input impedance. The error at 50% of pot travel is less than 5% with values from 10k to 100k.

The additional benefit of improved tracking does not apply to the active version (at least not to the same extent), so use the best pot you can afford to ensure accurate channel balance.

Better Balance Control (Contributed by Bernd Ludwig)

Bernd, a reader of The Audio Pages, has contributed a useful variation - in this case, a "better balance" control. Note that the configuration described requires a high impedance load, and the passive "Better Volume Control" cannot be used in this circuit. Used in the manner shown, it is a very similar concept to the better volume control of Figure 1, except it is (in a sense) the same idea in reverse.

Bear in mind that many (especially Japanese) designs use a specially designed pot for balance, and these are not suitable for the circuits shown below. These pots commonly have a centre detent, and the resistance of each track remains very low from the centre position to one end (or the other) of travel. These "special" pots are characterised by the level remaining constant in one channel or the other as the balance pot is moved. The overall law of these controls is (IMHO) unsatisfactory for hi-fi.

A standard configuration of Balance/Volume control using conventional pots (1 channel) is shown below:

Figure 4 - Conventional Balance / Volume Control

    BAL = 2,5 * VOL
    For example: VOL = 10k log, BAL = 25k lin

Adding a resistor R gives opportunity for two interesting improvements of the standard balance-volume-control networks. Note that the switch is optional, and may safely be left out.

Figure 5 - Improvement With Added Resistor

    A) R = VOL (for example, 10k)
The BAL-pot is 'virtually absent' when in centre position:

In centre position the resistive track of BAL only affects the load for the previous stage, since there is no current through the sliding contact (so you could open switch 'S' without changing anything at all - if you please). This seems to be reasonable: As long as you don't manipulate the balance control it is virtually absent from the circuit (no signal passes through its sliding contact). Hence quality (or age) of the BAL-pot doesn't matter at all then.

Sonic detriments can only come into play for two reasons:

  • If the resistive tracks of BAL are not absolutely symmetrical current through at least one of the sliding contacts will not be exactly zero at centre position (adding the switch 'S' would cure this entirely - but I doubt that there is any need for it).
  • If track resistance of a carbon pot (worst case scenario!) changes due to varying pressure of the sliding contact (induced by acoustical resonance, just like in the carbon microphones of veteran telephones), the load on the previous stage will change (but I suspect it might be really difficult to find a stage that will 'feel' it).
Thanks to R, the balance control operates conveniently slowly near the centre position and overall volume is affected significantly less than without it. This leads to another option:
    B) R = 4k7 (R = ~0.47*VOL)
The balance knob works without affecting the overall volume

This will give best operating convenience since the sound stage then moves from the left to the right without significant overall volume change. Input voltage on both channels constant and equal, the sum of left- and right-channel power remains approximately (+-0.2dB) constant across about 80% of the dial (which still works conveniently slowly about the centre position). I decided on the .47-factor after some PC-simulation and checked it by implementation in my preamp afterwards:

It works as expected indeed (there is just a slight increase of overall volume at the extreme right and left positions). I don't want to miss out on having a balance control any more, since there are in fact records which suffer from severe channel imbalance. Moving the armchair or the speakers is not a convenient cure for that. Moving the soloist two feet to the left or right without changing the overall volume, just by activating the balance knob, is the way to go.

Any compromise between 'golden-ear-' and 'maximum-convenience-' versions is possible by selecting a suitable 'R/Vol factor' between 1.0 and 0.47. :-).

The impedance of these "enhanced" networks is approximately that of 'VOL' alone (if R = Vol and BAL ~ 2 * VOL), so you can add BAL and R to any "purist's" design without changing critical parameters of the circuit (4-6dB attenuation by R will occur, of course, so you will have to add about 5 or 10 degrees of arc on the volume dial in future). Even when BAL is set to the extremes there is only a moderate change of load (max.: -30%) which will not upset any reasonable preamp.

If there already is a standard network in your amp, it is easy to add the Rs: Just solder them across the corresponding pins of the balance pot (on one channel from centre to the left and on the other from centre to the right!) The volume pot is not involved.

Electronic Night Light

This circuit for an electronic night light was submitted by Adam from Canada. I have provided the notes.

The two transistors are used as a direct coupled switch, Adam used 2SC711 but any general purpose transistor will do e.g. 2N3904, BC109C. The CDS photocell, type ORP12 is normally illuminated, therefore its resistance is low. The 50k control, the 1k resistor and the photocell form a potential divider which biases the first transistor. This transistor is on, its collector being held low, turns the last transistor and hence lamp and relay off.

In darkness, the resistance of the photocell becomes high and the first transistor switches off. The base voltage for the second transistor goes high, switching this transistor on and illuminating the lamp. Although Adam used a secondary supply of 3V , you could use any voltage and any lamp here. Make sure the relay contacts can handle the load. If using a large relay, it is preferable to wire a 1N4001 in reverse polarity across the coil. This will prevent the back EMF of the relay from damaging the transistors.

ECM Mic Preamplifier

A microphone amplifier that may be used with either Electret Condenser Microphone (ECM) inserts or dynamic inserts, made with discrete components.

Both transistors should be low noise types. In the original circuit, I used BC650C which is an ultra low noise device. These transistors are now hard to find but BC549C or BC109C are a good replacement. The circuit is self stabilizing and will set its quiescent point at roughly half the supply voltage at the emitter of Q2. This allows maximum output voltage swing and also the highest dynamic range.

The electret condenser microphone (ECM) contains a very sensitive microphone element and an internal FET preamp, a power supply in the range 2 to 10 volts DC is therefore necessary. Suitable ECM's may be obtained from Maplin Electronics. Although the schematic is drawn showing a three terminal ECM, two terminal ECM's may be used, the following page in the practical section shows the changes.

The 1k resistor limits the current to the mic. This resistor should be increased to 2k2 if a supply voltage above 12 Volts DC is used and is not needed if the Mic insert is dynamic. The first stage amplifier built around Q1 is run at a very low collector current. This factor contributes to a very high overall signal to noise ratio and low overall noise output. The emitter resistor of Q1 is decoupled by the 100u realizing a maximum gain for this stage. The noise response of the amplifier measured across the 10k load is shown below. Please note that this plot was made with the mic insert replaced by a signal generator.

The second stage, built around Q2 is direct coupled, this minimizes phase shift effects (introduced with capacitive and inductive coupling methods) and acheives a flat output response from 20Hz to over 100kHz. The frequency response measured across a 10k load resistor is plotted below simulated using a 12V power source:

The emitter voltage of Q2 is also fed back to the base of Q1 via resistive coupling. This also ensures bias stabilization againt temperature effects. Q2 operates in emitter follower mode, the voltage gain of this stage is less than unity, however, the overall voltage gain of the preamplifier is about 100x or 20dB as shown in the bode plot above. The output impedance is very low and well suited to driving cables over distances up to 50 meters. Screened cable therefore is not necessary.

This preamplifier has excellent dynamic range and can cope with anything from a whisper to a loud shout, however care should be taken to make sure that the auxiliary equipment i.e. amplifier or tape deck does not overload.

Doorphone Intercom

A simple 2 way Intercom based on the LM386386 using 8 ohm speakers.

For the first time, this circuit was designed by two authors, Mr Laurier Gendron of Burnaby in British Columbia, Canada, and myself. Please make sure you visit Laurier's web site, Handy Dandy Little Circuits. This page is also available in French by clicking on the flag.

In this doorphone circuit, an 8 ohm speaker is used both as a microphone and also an output device. The BC109C stage amplifies in common base mode, giving good voltage gain , whilst providing a low impedance input to match the speaker. Self DC bias is used allowing for variations in transistor current gain. An LM386 is used in non-inverting mode as a power amplifier to boost voltage gain and drive the 8 ohm speaker. The 10k potentiometer acts as the volume control, and overall gain may be adjusted using the 5k preset. The double pole double throw switch, reverses the loudspeaker positions, so that one is used to talk and the other to listen. Manually operating the switch (from inside the house) allows two way communication.

Digital Volume Control

This circuit could be used for replacing your manual volume control in a stereo amplifier. In this circuit, push-to-on switch S1 controls the forward (volume increase) operation of both channels while a similar switch S2 controls reverse (volume decrease) operation of both channels.

Digital Combination Lock

A multiple input combination loack using CMOS counter IC's. Flexibility and code change is allowed by changing output connections.

Notes The circuit above above makes use of the CMOS 4017 decade counter IC. Each depression of a switch steps the output through 0 - 9. By coupling the output via an AND gate to the next IC, a predefined code has to be input to create the output. Each PBS switch is debounced by two gates of a CMOS4001 quad 2-input NOR gate. This ensures a clean pulse to the input of each CMOS 4017 counter. Only when the correct number of presses at PBS A will allow PBS B to become active. This is similar for PBS C and PBS D. At IC4, PBS D must be pressed 7 times. Then PBS C is again pressed 7 times, stepping from output 1 to output 8. The AND gate formed around CMOS4081 then goes high, lighting the LED. The Reset switch can be pressed at any time. Power on reset is provided by the 100n capacitor near the reset switch. Below is a picture of one that I made about 15 years ago:

Unfortunately, this board was part of a much larger project containing multiple power supplies. One day whilst working on another circuit , I slipped with a wire and splashed 24 volts DC onto this board. There was a small spark, and puff of smoke before all this chips were cooked! If anyone does consider building such a circuit, then my advice would be to stop and look in your local electronic parts catalogue. There are now dedicated combination lock IC's with combinations many times greater than this circuit. Incidentally the numberof combinations offered here is 10 x 10 x 10 x 10 x 9 = 90,000.

Beetle Mk III Connect your Guitar to the airwaves

This project is based on our most stable FM transmitter.
It has a number of features, with a volume control to adjust the input level and a small, neat box to make it easier to attach to a guitar. The volume control is positioned at the end and when turned up fully, the transmitter provides fuzz (distortion).


Range: 20 - 50 metres
Supply: 3v
Current consumption: 7mA
Battery life: approx 50 hours
Tuning range: 80 -110MHz
(by stretching or compressing
the oscillator coil)
Fine tune by adjusting the air
trimmer (5MHz adjustment)
Stability: good.
Antenna length - 60 cm

The Beetle Mk III fitted into a PP12 box. All the
controls are mounted at the top, with the antenna
and guitar Jack at the bottom.
The wiring layout is in the article.

A close-up of the completed PCB

The Beetle MkIII Circuit


The circuit takes the signal from a guitar, radio or tape-recorder and injects it into an oscillator operating at 90MHz. This produces a carrier frequency - also called a Radio Frequency (RF).
This frequency is capable of being radiated to the surroundings and picked up by an FM radio.
But the signal produced by the oscillator stage is not very "powerful" so we pass it to an amplifying stage called a BUFFER or OUTPUT STAGE to increase the amplitude.
We can now connect an antenna to the output stage so the signal can be radiated.
The circuit can also be connected to an acoustic guitar by connecting a conducting microphone to the body of the guitar.
One of the best microphones for this purpose is a piezo diaphragm.
It is attached to the guitar with Blu-Tack and the two leads connected across the 100k input pot. The attenuation (volume control) can be adjusted to give the required signal level.

The Beetle MkIII kit also includes a battery holder


2 - 330R
1 - 470R
1 - 47k
1 - 150k
1 - 100k mini pot with shaft

2 - 10p ceramics
1 - 39p ceramic
1 - 1n ceramic
1 - 22n ceramics
1 - 100n mono-block (monolithic)
1 - Air trimmer 2p-10p

2 - BC 547 transistors
1 - 6 turn coil .5mm enamelled wire
1 - 5mm red LED
1 - LED mount
1 - 6.4mm mono plug
1 - SPDT mini toggle switch
1 - 30cm shielded wire
5 - 10cm hook-up wire
1 - 15cm tinned copper wire
1 - 30cm fine solder
2 - "N" cells
1 - "AAA" cell holder
1 - 60cm antenna wire
1 - PP12 box


Beetle MkIII Printed Circuit Board

Due to non-availability of "N" cell holders in Australia, we have had to include a AAA cell holder into the Kit, but with the use of a Hax saw the holder can be cut in two places as shown in the picture. The middle part is removed and the two ends jointed to become a "N" cell holder.
The first task in constructing the Beetle Mk III is to prepare the battery holder then Glue it in to the middle of the PP12 and leave to dry.

All the components fit on a PC board 14mm x 35mm. The pot, LED and volume control fit at one end of the box and are connected to the board with short leads.
The position of each component is marked on the PC board and you should have no difficulty constructing the project if you have made some of our other projects.
When all the parts have been fitted, three holes are drilled in one end of the box to take the switch, LED and volume control.

Tune across the FM and select a spot that is free of other transmissions, preferably at the low end of the band.
Switch the project ON and the LED will indicate power is applied.
Adjust the coil by expanding or compressing the turns until the radio goes silent.
This indicates the carrier from the transmitter is being picked up by the radio.
Now move the project away and fine tune the oscillator via the air trimmer to get the transmitter and radio on exactly the same frequency.
Fit the lid to the box and attach it to your guitar with Blu-Tack or Velcro. Insert the 6.4mm plug into the guitar and the project is complete.
Adjust the volume control to get the desired level. The volume can be increased to get a FUZZ effect.

Beetle MkIII is designed to transmit at about 88-92MHz with the 6 turn coil provided in the kit.
This is at the lower end of the commercial (88MHz to 108MHz) band. If you want to transmit below the band (86-88MHz) you will have to either compress the turns of the coil or add one more turn.
The best thing to do is add one more turn (there is enough wire provided in the kit for you to do this) so you can space the 7 turns neatly.
With a 7turn coil you will be able to go to 86MHz and tune the exact frequency you require with the aid of the air trimmer.
You must have a radio or tuner that will pick up the 86MHz band. If you don't, you can detune a radio by turning the trimmer capacitors on the back of the tuning gang so that the radio stations move up the dial and produce a space at the bottom of the band.
If you want to transmit above the commercial band, you can remove one of the turns of the coil and adjust the frequency with the trimmer capacitor. The transmitter may not have the same range at 108MHz as the transistors are operating at their maximum capability and the output power may be less at 108MHz.

The first thing to do is remove the 10p capacitor on the base of the output transistor and connect the LED Power Meter to the collector of the oscillator transistor.
At this point in time you should not have anything connected to the input of the project as it may upset the oscillator circuit. Also, the supply must be a set of 2 cells with short leads as anything else will also upset the performance of the circuit.
Turn the project ON and watch the needle on the multimeter. A reading will indicate the oscillator is working. The magnitude of the reading is not important at this stage, all we need to get is a reading.
The short lead to the LED Power Meter will act as an antenna so you can bring an FM radio near the project and tune across the dial to pick up the carrier. This will be detected as clean spot or "dead" spot on the dial. If this is not detected, the problem might be the oscillator is operating below the band. Try spreading the turns of the oscillator coil to raise the frequency.
If no reading is detected on the LED Power Meter, the oscillator section will be faulty. This consists of the oscillator transistor, 6 turn coil, air trimmer and 39p, 470R emitter resistor, 10p feedback capacitor. (make sure you have not removed the wrong 10p capacitor), 1n capacitor on the base, 47k base bias resistor and 22n across the power rails. Remove the output transistor and measure the supply current. It should be about 4 - 6mA.
If you think the trimmer has shorted, remove the coil and measure across the trimmer. Only if the reading is zero ohms, will the trimmer be damaged.
You cannot really measure the base or emitter voltage as any meter probe will act an antenna and draw off energy from the oscillator and prevent it from operating.
All you can do is replace the transistor and the 4 capacitors mentioned previously. Ceramic capacitors can go open circuit during soldering as the leads are soldered to the substrate and
this can crack or melt during soldering or when the component is fitted to the board - it all depends on the quality of the capacitor. If they are not damaged during soldering, they are a very reliable component.
This project has been used by some constructors as the basis of a very successful business, supplying FM transmitters to music groups.
Each transmitter can be tuned to a different frequency and mixed at the console. This gives the players freedom to move around the stage without tripping over wiring and leads.

Automatic Intruder Alarm

This is a simple single-zone burglar alarm circuit. Its features include automatic Exit and Entry delays and a timed Bell/Siren Cut-Off. It's designed to be used with the usual types of normally-closed input devices such as - magnetic reed contacts - micro switches - foil tape - and PIRs. But it can be Easily Modified to accept normally-open triggering devices - such as pressure mats.

Schematic Diagram:


It's easy to use. First check that the building is secure and that the green LED is lit. Then move SW1 to the "set" position. The red LED will light. You now have about 30 seconds to leave the building. When you return and open the door - the Buzzer will sound. You then have about 30 seconds to move SW1 to the "off" position. If you fail to do so - the relay will energize and the Siren will sound.

While at least one of the switches in the normally-closed loop remains open - the Siren will continue to sound. However, about 15-minutes after the loop has been restored - the relay will de-energize - the Siren will Cut-Off - and the alarm will Reset. Of course, you can turn the Siren off at any time by moving SW1 to the "off" position.

Because of manufacturing tolerances - the precise length of any delay depends on the characteristics of the actual components you've used in your circuit. But by altering the values of R3, R6 & R9 you can adjust the Exit, Entry and Bell Cut-Off times to suit your requirements. Increasing the values increases the time - and vice-versa.

The Support Material for this alarm includes a complete circuit description - a parts list - a step-by-step guide to construction - and more.

Veroboard Layout:

Audio Voice-Over Circuit

This is a circuit where a microphone and preamp circuit (voice circuit) have priority over any other audio signal. You can think of this as a one way intercom, if the main amplifier is used for listening to music, then when the push to talk switch is pressed, the amplifier is switched to the voice signal.

Application Notes:
In its simplest form, a voice-over unit is just a microphone and change-over switch feeding an amplifier, the output from the microphone having priority over the amplifiers audio signal when the "push-to-talk" switch is pressed. In this circuit, a preamplifier immediately follows the microphone and is designed to be used some distance away from the main amplifier. The changeover switch is nothing more than a relay with a single changeover contact. For completion, an amplifier based on the LM380 is shown. Three wires are needed to connect the remote microphone unit to the amplifier and switching unit.

Circuit Notes:
With reference to the above schematic, the two BC109C transistors are used to make a microphone preamplifier. The left hand BC109C operates in common emitter mode, the right hand emitter follower. The combination form a high gain, low output impedance amplifier, capable of driving a long audio cable. Screened cable is not required as the output impedance from the microphone pre-amp is very low, and will be immune to mains hum and background noise. The input is shown as a three wire Electret Condenser Microphone though two wire ECM's may also be used. The output of the pre-amp is via a 100uF capacitor and 1k resistor. The 1k resistor here plays an important role, eliminating the dc component of the audio output. (See also eliminating the DC "thump" also on this web site.) A cable of three or more wires is wired to the remote amplifier. The amplifier shown here is based on the National Semiconductor LM380. The input signal is passed via the normally closed contact of a changeover relay, the 10k potentiometer being the volume control for the audio input source. The 10k preset at the normally open contact allows volume control of the voice input, note that this signal has by-passed the normal volume control. At the remote end, when the push-to-talk switch is pressed, the relay will operate and the "voice" signal will be heard in the speaker. There will be no "thump" or "thud" on voice-over as direct current has been eliminated as already mentioned. A suitable application for this circuit would be for use in a remote location such as a workshop or shed.

Audio Signal Source

Audio Signal Source

This is a lovely project for home constructors keen on building their own test and measurement (T&M) equipment. It makes use of the ICL8038 signal generator chip, manufactured by Intersil. An improved version, made by Exar corp. is available (XR8038A). It can be used to produce three types of waveforms, sine, square and triangle. The frequency, amplitude and duty cycle can be varied, and selection of waveform is done digitally. To further reduce the complexity, a 3-to-1 switch may be used in place of the digital selection circuitry. I made use of the digital selection mechanism because switches available on the market are prone to dirt accumulation and poor contact quality. Besides, the digital method is a lot cooler!


The circuit shown below can be roughly divided into three parts: the oscillator based around the ICL8038 chip, the selection logic based on the CD4017 and CD4066 and the offset generation and output buffers, based on the LF412. Apologies for the cramped schematic, I had to keep the image size small, and the width under 640 pixels!
The oscillator is a standard 8038-based oscillator circuit, taken from the ICL8038 datasheet. The timing resistor chosen is rather small, to give a wide range of frequencies. This range might be a little too large, making precise frequency setting difficult. In that case, the freqency range may be split into two parts, using two capacitors which can be switched using an SPDT switch. Note that the 8038 is powered from a split supply, not a single supply, to generate a symmetrical waveform without the need for capacitor coupling. Two sine wave adjustment terminals (Pins 1 and 12) are provided, however only one is used. This gives a sinewave distortion of about 1%. To achieve better distortion figures, the circuit shown in Figure 4 of the ICL8038 datasheet may be used. The 8038 is powered from slightly less than +8V to allow the tuning voltage to go above the supply rail. This allows for maximum sweep range (1000:1), however the output waveform tends to be slightly asymmetric because of this. This may be compensated using the offset control R10. R2 controls the duty cycle of the oscillator. R7 acts as the sinewave distortion adjustment. The square wave output of the 8038 is an open-collector output. Hence, a 1k pullup resistor is provided. The sine and triangle outputs are about 5Vpp, while the square wave is 16Vpp. Hence to equalize the different outputs, the square wave is attenuated using a fixed attenuator formed by R6 and R9. A 47k pot may be substituted to make the attenuation level adjustable.

Audio Signal Source schematic
One of the three outputs are then selected using the digital selection circuit. This is based on the CD4066 quad analog bilateral switch and the CD4017 decade counter with fully decoded outputs. I havent used the 4051-series Analog MUXes here because getting a counter to drive this is a bit of a pain. The 4066 essentially acts like four switches under electronic control. The voltages switched must not be above or below the supply voltages. Hence, the 4066 is given both +8 and -8. Now, however, the selection inputs to the 4066 cannot be with reference to ground, and since the selection inputs are driven by the 4017, it too must have a split supply. Hence, note that both chips have ground connected to -8V and not 0V. The 4017 is wired so that it resets itself when the fifth state is selected. The clock to the 4017 comes from a pushbutton switch with a pullup to +8. A 0.1μ capacitor across the switch serves as a contact debouncer. This prevents multiple triggering of the 4017 from a single keypress. Hence, each output is successively chosen when the switch is pressed. The fourth state corresponds to the output being off (DC). On the fifth press, the counter resets itself and selects sine wave again.
The final stage is the offset adjust, amplitude adjust and output driver stage. The offset adjust is an inverting amplifier whose reference pin is not at ground. It is instead attached to a 1k pot's wiper, connected across the supply rails. Hence, when the reference pin is taken off ground, the amplifier introduces a DC shift corresponding to the product of the gain and the reference voltage. The gain of this amplifier is chosen to be about 1.5, to raise the amplitude of the sine and triange waves (but not high enough to cause distortion). The output is passed through an attenuator R12 before going to the output driver voltage follower. This stage uses an LF412 opamp, since the risetime of the square wave imposes a high slew rate on the opamp. Cheaper opamps such as the LM358 have poor slew rates compared to the LF412. Also, the LF412 has a robust output stage. A mistake in this design is putting the attenuator after the stage introducing a voltage offset. This means that even the offset is attenuated along with the signal... not desireable behaviour. The solution would either require a pot with a high resistance (100k or more) in place of the 10k unit here, connected before the offset-introducing inverting amplifier U3B. No pot is required between U3B and U3A.

Audio Signal Source Power Supply
The power supply is a regular split-supply design based on a 78L08 and 79L08 linear regulators. I made use of a 9-0-9 transformer, which was a bit risky (very low headroom voltage for the regulators), but then my dealer didnt have a stock of 12-0-12 transformers. The output of the transformer is rectified and smoothed using 100μF capactors. The float-voltage measured here for the 9-0-9 transformer was +/- 13V. The unregulated DC is regulated by the 78L08 and 79L08 to give the regulated supply rails for the circuit.


Again, the usual breadboard/veroboard. I used Relimate connectors for all the connections to and from the board. As a retrofit, I also added a connector to make available the +/- 8V rails to outside projects, since I do not own a bench split supply. The whole works was shoved into a small plastic chocolate box (!!!), but I haven't got down to drilling holes for the pots, switch, etc. etc. Performance was quite good, I measured frequencies up to 60kHz, with a rather clean looking sine wave. The square wave was attenuated too much, but that was OK by me. The rise time for the square wave was very good. As a test, replacing the LF412 with an LM358 resulted in very poor square wave output (it almost looked like a sine!), and other waveforms seemed to have crossover distortion (*shrug*... crossover distortion in an opamp!!!) Amplitude could be adjusted from 0V to about 7Vpp for all waveforms. Contact bounce was still a bit of a problem for the counter, sometimes the counter moved two places rather than one. A higher value capacitor will help a bit. I attached three LEDs through 1k resistors to each of the three used outputs of the 4017 to indicate which waveform is selected. This wasnt included in the schematic due to space constraints.
It would be wise to go through the excellently written "Everything You Always Wanted to Know About the ICL8038" (AN013.1) by Intersil. It is available at Intersil's website, along with the ICL8038 datasheet.


As I mentioned, move the attenuator to before the offset stage. I'm planning to make a frequency synthesizer based on a similar circuit with a PLL, using the 8038 as the VCO. It should be under computer control or use a microcontroller to form a stand-alone instrument.

Amp with Tone Controls & Soft Switching

Built around an LM380, this amplifier includes tone controls and electronic "soft switching". The soft switching circuitry ensures power is built up gradually eliminating the dc thump.

The soft switching is enabled by a BD131 transistor wired as a switch in emitter follower configuration. The collector is wired to a permanent supply voltage, the 2H series inductor serves only to filter out power supply hum. This inductor is not too important and may be omitted if the DC supply is adequately smoothed. The control voltage is applied to the BD131 base terminal, the 10u capacitor and 10k resistor having a dual purpose:-
i) a gradual charge of the 10u capacitor ensures that the transistor will switch linearly from 0 volts to full supply, and
ii) serves as a hum filter ensuring a very smooth dc supply to the amplifier and tone controls.
LED1 will light when the amplifier is on. The control voltage should ideally be 0 volts when the amplifier is off and full supply voltage when on. The LM380 is shown driving two 8 ohm loupspeakers, the load is therefore 4 ohms. The 4u7 capacitor acts as a crude crossover, lower frequencies are impeded and so this loudspeaker may be a "tweeter" type.

Tone Controls:
The input of this is amplifier is via a tone control based on the baxendall design. The first BC109C serves as a buffer, offering a high input impedance. The output signal fed via a 10u capacitors reaches the tone control network. This passive network of resistors and capacitors attenuates high and low frequencies. The bass control is centered on 100Hz and treble control 10kHz. The second BC109C amplifies the losses from the tone control, overall the tone control provides roughly +6dB boost and -20dB cut. See the bode plot below:

Bode response of Tone Controls:

The traces show maximum lift, maximum cut and the response with tone controls in the centre position. There may be difficulty in obtaining BC109C transistors, but BC108C transistors will make an ideal replacement.

LM380 Pinout:

More IC Pinouts may be found on my IC pinout page in the Practical section.

AC Switch

An AC triggered switch for low frequency signals.

This is a basic ac voltage operated switch made from discrete components. Both Q1 and Q2 work as common emitter amplifiers, but the biasing of Q1 is arranged by R3 and R4 so that about 0.5V is applied to its base; so with no input signal both transistors and the load is off.

With an input signal greater than 0.3V pk-pk (100mV RMS) the positive half of the waveform will switch on Q1, and Q2 and the relay. As the input signal switches to its negative transition, Q1 will switch off, but base current in Q2 continues to flow via C2, so Q2 and hence relay load remain on. This will happen for any ac signal within 50 to 1000Hz. R1 prevents excessive base current flowing in Q2, if required a series reistor of 100 ohms can be included with C1 to reduce excessive current flow, though this may decrease sensitivity.

C2 has a dual purpose; as well as smoothing the input signal, it adds a delay to the on/off operation. The delay is dependent on the value of C2 and the coil resistance of the relay. Instead of a relay, a LED and series resistor of 1k could be used instead, however the relay has the advantage of being able to switch large loads on and off.

A Thermostat With Adjustable Hysteresis

Circuit Notes
A thermostat doesn't try to maintain a constant temperature. In order to do so - it would have to keep switching on and off every few seconds. Instead - it keeps the temperature within a specific range. When the preset temperature has been reached - it switches off. And it only switches on again - when there has been a significant change in temperature.

The difference between the temperature at which the thermostat switches off - and the temperature at which it switches on again - is the hysteresis. Without this hysteresis - your central heating, refrigerator etc. would keep switching on and off every few seconds.

This particular circuit energizes the relay when the temperature falls - and de-energizes the relay when the temperature rises again. If you replace the pnp transistor (BC557) with an npn transistor (BC547) - the circuit will operate the other way round.

In order to minimize power consumption - choose the configuration that energizes the relay for the shorter time period. If it's going to be hot most of the time - choose the one that energizes the relay when the temperature falls (BC557). If it's going to be cold most of the time - choose the one that energizes the relay when the temperature rises (BC547).

Temperature Range
There are many factors that influence how a thermostat will perform. For example - heat energy stored in a heater may go on raising the temperature - even after the heater has switched off. Similarly - even after the heater has switched on again - the temperature may continue to fall - while the cold heater warms itself up. In other words - the system you're trying to control - may have some hysteresis of its own.

The temperature at which the relay will energize - is controlled by the voltage on pins 5 & 6. And the temperature at which the relay will de-energize - is controlled by the voltage on pins 1 & 2. The difference between the two temperatures - the hysteresis - is controlled by the value of R3.

With the component values used in the prototype - the temperature at which the relay energizes can be adjusted - from roughly 22°C (°72F) to 29°C (84°F). And the hysteresis is around 4°C (7°F). Because of manufacturing tolerances - your results are likely to be different. But - by changing the values of R1, R2 & R3 - you can select both the temperature range available - and the amount of hysteresis.

Thermostat with Adjustable Hysteresis Prototype

The value of R1 sets the width of adjustment available. The higher the value of R1 - the wider the range of possible temperature settings. However - if you make the range of adjustment too wide - setting a precise temperature becomes more difficult. A lower value pot - or a multi-turn pot - will make fine adjustment much easier. Turning R1 to the right increases the temperature setting. And turning R1 to the left reduces the temperature setting.

R2 lets you choose the area of the temperature scale in which the thermostat is to operate. Reducing the value of R2 gives access to higher temperatures. Increasing the value of R2 gives access to lower temperatures. Use R2 to take you close to your target temperature - and use R1 to provide the fine adjustment.

R3 controls the amount of hysteresis. Increasing the value of R3 increases the hysteresis - and decreasing the value of R3 reduces the hysteresis. The actual amount of hysteresis provided by any given value resistor - depends on the area of the temperature scale you've selected. The resistance required - per degree of hysteresis - is higher at low temperatures - and lower at high temperatures.

Component tolerances make it difficult to predict the precise results any set of resistor values will give. However, the Support Material for this circuit includes a spreadsheet that will help you choose the right value resistors for your application. It also includes a parts list - a step-by-step guide to the construction of the circuit board - a detailed circuit description - a photo of the prototype - and more.

Do not use the "on-board" relay to switch mains voltage. The board's layout does not offer sufficient isolation between the relay contacts and the low-voltage components. If you want to switch mains voltage - mount a suitably rated relay somewhere safe - Away From The Board. Alternatively - you may be able to drive the mains device directly via an Optical Isolator - see below.

Veroboard Layout

The Temperature Sensor
The mass of the thermistor is small - and it will react quickly to changes in temperature. This makes it sensitive to thermal currents. One solution is to create an enclosed environment - where the thermistor is protected from momentary draughts - but can still respond to slower changes in the overall ambient temperature. Another solution is to increase its mass - by attaching it to a small metal heatsink.

Setting the operating temperature is best carried out "in situ" using trial-and-error. If the relay reacts at too low a temperature - turn R1 a little to the right. If it reacts at too high a temperature - turn R1 a little to the left. If you use a multi-turn pot - you can make very fine adjustments.

The Supply Voltage
The circuit is designed for a 12-volt power supply. However - it will work at anything from 5 to 15-volts. All you need do is select a relay with a coil voltage that suits your supply. And make sure that the coil doesn't draw more than about 50mA - otherwise the transistor might be overloaded. I've used a single-pole relay in the diagrams - but you can use a multi-pole relay if it suits your application. The Support Material for this circuit includes a parts list - a step-by-step guide to the construction of the circuit board - a detailed circuit description - a photograph of the prototype - and more.

Alternative Optical-Isolator Output

If you intend to use the circuit to control mains devices - the relay can be replaced by a whole range of optical-isolators - such as the MOC 3021. Please note that I am unable to help with the selection of suitable types of isolator - and I cannot supply circuit diagrams or connection details.

300W/500 Subwoofer Power Amplifier


There are some important updates to this project, as shown below. Recent testing has shown that with the new ON Semi transistors it is possible to obtain a lot more power than previously. The original design was very conservative, and was initially intended to use 2SA1492 and 2SC3856 transistors (rated at 130W) - with 200W (or 230W) devices, some of the original comments and warnings have been amended to suit.

30 Jul 2003 - OnSemi has just released a new range of transistors, designed specifically for audio applications. These new transistors have been tested in the P68, and give excellent results. As a result, all previous recommendations for output transistors are superseded, and the new transistors should be used.

The output devices are MJL4281A (NPN) and MJL4302A (PNP), and feature high bandwidth, excellent SOA (safe operating area), high linearity and high gain. Driver transistors are MJE15034 (NPN) and MJE15035 (PNP). All devices are rated at 350V, with the power transistors having a 230W dissipation and the drivers are 50W.

23 Sept 2003 - The new driver transistors (MJE15034/35) seem to be virtually impossible to obtain - ON Semi still has no listing for them on the website. The existing devices (well known and more than adequate) are MJE15032 (NPN) and MJE15033 (PNP), and these will substitute with no problems at all. It is also possible to use MJE340 and MJE350 as originally specified (note that the pinouts are reversed between the TO-126 and TO-220 devices).

Note that some component values have been changed! The layout is the same, but the changes shown will reduce dissipation in Q7 and Q8 under light load conditions.

Having built a couple of P68 amps using these transistors, I recommend them highly - the amplifier is most certainly at its very best with the high gain and linearity afforded by these devices. Note that there are a few minor changes to the circuit (shown below).

With ±70V supplies, the input and current source transistors must be MPSA42 or similar - the original devices shown will fail at that voltage! Note that the MPSA42 pinout is different from the BC546s originally specified. Full details of transistor pinouts are shown in the construction article (available to PCB purchasers only).

High power amps are not too common as projects, since they are by their nature normally difficult to build, and are expensive. A small error during assembly means that you start again - this can get very costly. I recommend that you use the PCB for this amplifier, as it will save you much grief. This is not an amp for beginners working with Veroboard!

The amplifier can be assembled by a reasonably experienced hobbyist in about three hours. The metalwork will take somewhat longer, and this is especially true for the high continuous power variant. Even so, it is simple to build, compact, relatively inexpensive, and provides a level of performance that will satisfy most requirements.


  • This amplifier is not trivial, despite its small size and apparent simplicity. The total DC is over 110V (or as much as 140V DC!), and can kill you.
  • The power dissipated is such that great care is needed with transistor mounting.
  • The single board P68 is capable of full power duty into 4 Ohm loads, but only at the lower supply voltage.
  • For operation at the higher supply voltage, you must use the dual board version.
  • There is NO SHORT CIRCUIT PROTECTION. The amp is designed to be used within a subwoofer or other speaker enclosure, so this has not been included. A short on the output will destroy the amplifier.


Please note that the specification for this amp has been upgraded, and it is now recommended for continuous high power into 4 Ohms, but You will need to go to extremes with the heatsink (fan cooling is highly recommended). It was originally intended for "light" intermittent duty, suitable for an equalised subwoofer system (for example using the ELF principle - see the Project Page for the info on this circuit). Where continuous high power is required, another 4 output transistors are recommended, wired in the same way as Q9, Q10, Q11 and Q12, and using 0.33 ohm emitter resistors.

Continuous power into 8 ohms is typically over 150W (250W for ±70V supplies), and it can be used without additional transistors at full power into an 8 ohm load all day, every day. The additional transistors are only needed if you want to do the same thing into 4 ohms at maximum supply voltage! Do not even think about using supplies over ±70V, and don't bother asking me if it is ok - it isn't!

The circuit is shown in Figure 1, and it is a reasonably conventional design. Connections are provided for the Internal SIM (published elsewhere on the Project Pages), and filtering is provided for RF protection (R1, C2). The input is via a 4.7uF bipolar cap, as this provides lots of capacitance in a small size. Because of the impedance, little or no degradation of sound will be apparent. A polyester cap may be used if you prefer - 1uF with the nominal 22k input impedance will give a -3dB frequency of 7.2Hz, which is quite low enough for any sub.

Figure 1 - Basic Amplifier Schematic

The input stage is a conventional long-tailed pair, and uses a current sink (Q1) in the emitter circuit. I elected to use a current sink here to ensure that the amp would stabilise quickly upon application (and removal) of power, to eliminate the dreaded turn on "thump". The amp is actually at reasonably stable operating conditions with as little as +/-5 volts! Note also that there are connections for the SIM (Sound Impairment Monitor), which will indicate clipping better than any conventional clipping indicator circuit. See the Project Pages for details on making a SIM circuit. If you feel that you don't need the SIM, omit R4 and R15.

The Class-A driver is again conventional, and uses a Miller stabilisation cap. This component should be either a 500V ceramic or a polystyrene device for best linearity. The collector load uses the bootstrap principle rather than an active current sink, as this is cheaper and very reliable (besides, I like the bootstrap principle :-)

All three driver transistors (Q4, 5 & 6)must be on a heatsink, and D2 and D3 should be in good thermal contact with the driver heatsink. Neglect to do this and the result will be thermal runaway, and the amp will fail. For some reason, the last statement seems to cause some people confusion - look at the photo below, and you will see the small heatsink, 3 driver transistors, and a white "blob" (just to the left of the electrolytic capacitor), which is the two diodes pressed against the heatsink with thermal grease.

C11 does not exist on this schematic, so don't bother looking for it. It was "mislaid" when the schematic was prepared, and I didn't notice until someone asked me where and what it was supposed to be. Sorry about that.

It is in the output stage that the power capability of this amp is revealed. The main output is similar to many of my other designs, but with a higher value than normal for the "emitter" resistors (R16, R17). The voltage across these resistors is then used to provide base current for the main output devices, which operate in full Class-B. In some respects, this is a "poor-man's" version of the famous Quad current dumping circuit, but without the refinements, and in principle is the same as was used in the equally famous Crown DC300A power amps.

Although I have shown MJL4281A and MJL4302A output transistors, because they are new most constructors will find that these are not as easy to get as they should be. The alternatives are MJL3281/ MJL1302 or MJL21193/ MJL21194.

Note: It is no longer possible to recommend any Toshiba transistors, since they are the most commonly counterfeited of all. The 2SA1302 and 2SC3281 are now obsolete - if you do find them, they are almost certainly fakes, since Toshiba has not made these devices since around 1999~2000.

Use a standard green LED. Do not use high brightness or other colours, as they may have a slighty different forward voltage, and this will change the current sink's operation - this may be a miniature type if desired. The resistors are all 1/4W (preferably metal film), except for R10, R11 and R22, which are 1W carbon film types. All low value resistors (3.3 ohm and 0.33 ohm) are 5W wirewound types.

Because this amp operates in "pure" Class-B (something of a contradiction of terms, I think), the high frequency distortion will be relatively high, and is probably unsuited to high power hi-fi. At the low frequency end of the spectrum, there is lots of negative feedback, and distortion is actually rather good, at about 0.04% up to 1kHz. My initial tests and reports from others indicate that there are no audible artefacts at high frequencies, but the recommendation remains.

Power Dissipation Considerations
I have made a lot of noise about not using this amp at ±70V into 4 ohms without the extra transistors. A quick calculation reveals that when operated like this, the worst case peak dissipation into a resistive load is 306W (4Ω/ ±70V supplies). The four final transistors do most of the work, with Q7 and Q8 having a relatively restful time (this was the design goal originally). Peak dissipation in the 8 output devices is around 70W each.

Since I like to be conservative, I will assume that Q7 and Q8 in the updated schematic shown contribute a little under 1A peak (which is about right). This means that their peak dissipation is around 18W, with the main O/P devices dissipating a peak of 70W each. The specified transistors are 230W, and the alternatives are 200W, so why are the extra transistors needed?

The problem is simple - the rated dissipation for a transistor is with a case temperature of 25°C. As the amp is used, each internal transistor die gets hot, as does the transistor case - the standard derating curves must be applied. Add to this the reactive component as the loudspeaker drives current back into the amp (doubling the peak dissipation), and it becomes all too easy to exceed the device limits. The only way that this amp can be used for continuous high power duty with ±70V supplies and a 4Ω loudspeaker load is to keep the working temperature down to the absolute minimum - that means four output devices per side, a big heatsink and a fan!

Figure 1a - Double Output Stage

Figure 1A shows the doubled output stage, with Q9, Q10, Q11 and Q12 simply repeated - along with the emitter resistors. Each 1/2 stage has its own zobel network and bypass caps as shown, as this is the arrangement if the dual PCB version is built. When you have this many power transistors, the amp will happily drive a 4 ohm load all day from ±70V - with a big enough heatsink, and forced cooling. Over 500W is available, more than enough to cause meltdown in many speakers!

A Few Specs and Measurements
The following figures are all relative to an output power of 225W into 4 ohms, or 30V RMS at 1kHz, unless otherwise stated. Noise and distortion figures are unweighted, and are measured at full bandwidth. Measurements were taken using a 300VA transformer, with 6,800uF filter caps.

Mains voltage was about 4% low when I did the tests, so power output will normally be slightly higher than shown here if the mains are at the correct nominal voltage. Figures shown are measured with ±56V nominal, with the figure in (brackets) estimated for ±70V supplies.

Voltage Gain27dB27dB
Power (Continuous)153W (240W)240W (470W)
Peak Power - 10 ms185W (250W)344W (512W)
Peak Power - 5 ms185W (272W)370W (540W)
Input Voltage1.3V (2.0V) RMS1.3V (2.0V) RMS
Noise *-63dBV (ref. 1V)-63dBV (ref. 1V)
S/N Ratio *92dB92dB
Distortion (@ 4W)0.04% (1 Khz)0.04% (1 Khz)
Distortion (@ 4W)0.07% (10 kHz)0.07% (10 kHz)
Slew Rate> 3V/us> 3V/us

* Unweighted

These figures are quite respectable, especially considering the design intent for this amp. While (IMO) it would not be really suitable for normal hi-fi, even there it is doubtful that any deficiencies would be readily apparent, except perhaps at frequencies above 10kHz. While the amp is certainly fast enough (and yes, 3V/us actually is fast enough - response extends to at least 30kHz, but not at full power), the distortion may be a bit too high.

Note that the "peak power" ratings represent the maximum power before the filter caps discharge and the supply voltage collapses. I measured these at 5 milliseconds and 10 milliseconds. Performance into 4 ohm loads is not quite as good, as the caps discharge faster. The supply voltage with zero power measured exactly 56V, and collapsed to 50.7V at full power into 8 ohms, and 47.5V at full power into 4 ohms.

The photo does not show the silk screened component overlay, since this is the prototype board. The final boards have the overlay (as do all my other boards). The observant reader will also see that the 5W resistor values are different from those recommended - this was an early prototype using 130W transistors.

As can be seen, this is the single board version. The driver transistors are in a row, so that a single sheet aluminium heatsink can be used for all three. Holes are provided on the board so the driver heatsink can be mounted firmly, to prevent the transistor leads breaking due to vibration. This is especially important if the amp is used for a powered subwoofer, but will probably not be needed for a chassis mounted system.

The driver and main heatsinks shown are adequate for up to 200W into 4 ohms with normal program material. The power transistors are all mounted underneath the board, and the mounting screw heads can be seen on the top of the board.

Deceptively simple, isn't it?

Power Supply

WARNING: Mains wiring must be performed by a qualified electrician - Do not attempt the power supply unless suitably qualified. Faulty or incorrect mains wiring may result in death or serious injury.

The basic power supply is shown in Figure 2. It is completely conventional in all respects. Use a 40-0-40 V transformer, rated at 300VA for normal use. For maximum continuous power, a 50-0-50V (500VA or more) transformer will be needed. This will give a continuous power of about 450W, and peak power of over 500W is possible with a good transformer. Remember my warnings about using the amp in this way, and the need for the additional output transistors, big heatsink and fan.

Figure 2 - Basic Power Supply Circuit

For 115V countries, the fuse should be 6A, and in all cases a slow blow fuse is required because of the inrush current of the transformer. For anything above 300VA, a soft-start circuit is highly recommended (see Project 39).

The supply voltage can be expected to be higher than that quoted at no load, and less at full load. This is entirely normal, and is due to the regulation of the transformer. In some cases, it will not be possible to obtain the rated power if the transformer is not adequately rated.

Bridge rectifiers should be 35A types, and filter capacitors must be rated at a minimum of 63V (or 75V if you use 70V supplies). Wiring needs to be heavy gauge, and the DC must be taken from the capacitors - not from the bridge rectifier.

Although shown with 4,700uF filter capacitors, larger ones may be used. Anything beyond 10,000uF is too expensive, and will not improve performance to any worthwhile degree. Probably the best is to use two 4,700uF caps per side (four in all). This will actually work better than a single 10,000uF device, and will be cheaper as well.

NOTE: It is essential that fuses are used for the power supply. While they will not stop the amp from failing (no fuse ever does), they will prevent catastrophic damage that would result from not protecting the circuit from over-current conditions. Fuses can be mounted in fuseholders or can be inline types. The latter are preferred, as the supply leads can be kept as short as possible. Access from outside the chassis is not needed - if the fuses blow, the amplifier is almost certainly damaged.